Signal reception method and device

ABSTRACT

The invention concerns a method and a device for receiving a signal transmitted by a source by means of a plurality of antennas, in which a plurality of beam signals are obtained by weighting the signals received by the different antennas. The beam signals are correlated with the replica of a reference signal transmitted by the said source and then combined by means of coefficients obtained from correlation results.

[0001] The present invention concerns in general terms a method anddevice for receiving signals by means of an array of antennas. Itapplies particularly to the reception by a base station of signalstransmitted by one or more mobile terminals.

[0002] Beamformig is well known in the held of narrow band antennaprocessing. It uses an array of antennas, generally linear and uniform(that is to say with a constant spacing between the antennas) and asignal weighting module. In order to form a beam in reception mode, thesignals received by the different antennas in the array are weighted bya set of complex coefficients before being added.

[0003] If the vector of the signals received by the N antennas in thearray is denoted x=(x₀,x₁, . . . ,x_(N-1))^(T) and the vector of theweighting coefficients (which will be referred to more simply as theweighting vector) is denoted w=(w₀,w₁, . . . ,w_(N-2))^(T), the outputsignal y of the beamformer will be written:

y=w ^(T) x  (1)

[0004] When it is wished to receive the signal transmitted by a givensource (for example a mobile terminal), the weighting coefficients w_(i)are determined so that the reception beam points in the direction ofthis source. In the majority of cases, the direction of arrival (DOA) ofthe signal is not known and use is made of one of a number of estimationmethods available in the state of the art (for example MUSIC, ESPRIT andderivatives thereof).

[0005] If a priori knowledge of a reference signal transmitted by thesource (or an estimation of a transmitted signal) is available, it ispossible to determine the weighting coefficient so as to minimise theroot mean square error between the output of the beamformer and thereference signal. Equation (1) represents a spatial filtering operationand the coefficients of the optimum filter can then be obtained by meansof the Wiener-Hopf equation:

w ^(T) =R _(dx) R _(xx) ⁻¹  (2)

[0006] where R_(xx) is the autocorrelation matrix of the signalsreceived, that is to say R_(xx)=E(xx^(H)), and R is the matrix (in thiscase, here, that of a linear form) for correlation of the referencesignal d with the received signals, that is to say R_(dx)=E(dx^(H)).These matrices must be updated when the spatial transfer functionvaries.

[0007] If it is wished to receive the signals transmitted by a pluralityof sources S₀, . . . S_(M−1) by means of the array of antennas, itsuffices to form a plurality M of beams pointing in the respectivedirections of arrival of the signals transmitted by the differentsources. If a priori knowledge of the reference signals d₀, . . .,d_(M−1) transmitted by the different sources is available, it ispossible to determine the weighting coefficients of each of the Mbeamformers by the matrix:

W ^(T) =R _(dx) R _(xx) ⁻¹  (3)

[0008] where R_(xx) is the autocorrelation matrix of the signalsreceived, that is to say d=(d₀, . . . ,d_(M−1))^(T) is the vector of thereference signals and R_(dx) is the correlation matrix between thereference signals and the received signals, that is to sayR_(dx)=E(dx^(H)). The respective weighting vectors of the differentbeamformers are given by the columns of the matrix W.

[0009] This technique has in particular been applied in the field ofmobile telecommnunications, notably to the CDMA (Code Division MultipleAccess) systems where it is known by the acronym CAAAD (CoherentAdaptive Array Diversity). A CAAAD receiver is illustrated schematicallyin FIG. 1. It comprises an array of antennas 100 ₀, . . . ,100 _(N−1),beamformers 110 ₀, . . . ,110 _(M−1) receiving the N antennas signalsand supplying the beam signals y₀, . . . ,y_(M−1) to rake receivers 120₀, . . . ,120 _(M−1). Each of the M beamformers points towards thedirection of arrival of the signal transmitted by mobile terminal. Atthe output of each beamformer, a RAKE receiver effects an MRC (MaximumRatio Combining) combination of the signals relating to the differentpropagation paths between the mobile terminal in question and the arrayof antennas. It is assumed that all the propagation paths are receivedby the beamformer.

[0010] The CAAAD reception technique is optimum according to a criterionfor minimisation of the root mean square error (MMSE or Minimum MeanSquare Error) but is difficult to implement. This is because it requiresas many beamformers as there are mobile terminals in the cell. Inaddition, the inversion of the matrix R_(xx) (or equivalently theresolution of a system of N linear equations with N unknowns) is acomplex operation which, in practice, will have to be performed bydedicated circuits. Furthermore, since the beamforming has to beadaptive in order to follow the movement of the terminal, this operationmust be performed frequently, which burdens the calculation resources ofthe receiver.

[0011] One reception technique, more robust and of lower performance,consists of forming a large number of fixed beams, for example beamswhich are angularly equally distributed, and, for each mobile terminal,selecting the one which supplies the signal with the highest powercoming from the said terminal. A receiver operating according to thisprinciple is illustrated in FIG. 2. The beamformers 210 ₀, . . . ,210_(L−1) with L>>N, where N is the number of mobile terminals, form Lfixed beams from signals received by the antennas 200 ₀, . . . ,200_(N−1). The beam signals y₀, . . . ,y_(L−1) output from the beamformersare transmitted to a plurality of selection modules 215 ₀, . . ., 215_(M−1), each selection module being associated with a mobile terminal.Decision modules 216 ₀, . . . ,216 _(M−1) supply to each of theselection modules the index of the beam to be selected. A decisionmodule 216 _(i) associated with a terminal i ∈{0, . . . ,N−1 } effects acorrelation between the beam signals y₀, . . . ,y_(L−1) and a referencesignal transmitted by this terminal, chooses the index of the beamsupplying the highest energy and transmits it to the selection module215 _(i). The reference signals of the different terminals must bechosen so as to be orthogonal: for a CDMA systems use would be made ofthe pilot symbols transmitted over the uplink channel DPCCH (DedicatedPhysical Control CHannel). The beam signal selected by the selectionmodules 215 ₀, . . . ,215 _(M−1) are then supplied to RAKE receivers 222₀, . . . ,220 _(M−1) exploiting the reception diversity within theselected fixed beam.

[0012] This reception technique does however have many drawbacks. Itrequires on the one hand the formation of a large number of beams inorder to obtain a good angular aiming resolution. Moreover, it functionsbadly in a micro or pico-cellular environment in which signal propagatesalong paths which are broadly angularly scattered because of multiplereflections. In this case, the fixed beam selected for a terminal willvery often contain only the propagation path with the highest power andthe other paths will not be used by the RAKE receiver downstream The aimof the invention is to propose a reception method by means of an arrayof antennas which does not have the aforementioned drawbacks, inparticular which allows optimum reception of signals transmitted by aplurality of sources without requiring large calculation resources.

[0013] To this end, the invention is defined by a method of receiving asignal transmitted by a source by means of an array of antennas, inwhich a plurality of beam signals are formed by weighting of the signalsreceived by the different antennas. According to this method each of thesaid beam signals is correlated with the replica of a reference signaltransmitted by the said source and the said beam signals are combined bymeans of coefficients obtained from the correlation results.

[0014] The coefficients can be obtained by complex conjugation of thesaid correlation results and the beam signals can be normed prior totheir combination.

[0015] Typically, the coefficients are obtained from a linear functionof the previously conjugated correlation results. Advantageously, thecoefficients are obtained by the matrix operation γ=Ω⁻¹α* where γ=(γ₀, .. . ,γ_(L−1))^(T) is the vector of the said coefficients, α=(α₀, . . .,α_(L−1))^(T) is the vector of the said correlation results, L is thenumber of beams and Ω is the matrix of size L×L whose coefficients are afunction of the correlation of the gain functions relating to thedifferent beams.

[0016] According to a first variant, the coefficients Ω_(l′l) areobtained by Ω_(l′l)=g_(l′) ^(H)g_(l) where g_(l) and g_(l′) are vectorsformed by angular sampling of the gain functions relating to the beamsof indices l and l′.

[0017] According to a second variant, the coefficients Ω_(l′l) areobtained by Ω_(l′l)=∫G_(l′)(θ)*G_(l)(θ)dθ where G_(l)(θ) and G_(l′)(θ)are the gain functions relating to the channel of indices l and l′respectively and where the summing is effected over the angular scanningrange defined by the array.

[0018] Advantageously, the signal resulting from the combination of thesaid beam signals is subjected to a putting back in phase of thecomponents relating to the different propagation paths between the saidsource and the said array.

[0019] The invention is also defined by a method of receiving, by meansof an array of antennas, signals transmitted by a plurality of sources,in which a plurality of beam signals are formed by weighting of thesignals received by the different antennas and in which, for eachsource, the said beam signals are correlated with a replica of thereference signal transmitted by the said source and the said beamsignals are combined by means of coefficients obtained from thecorrelation results.

[0020] Advantageously, prior to the combination step and for each beamsignal, the components relating to the different propagation pathsbetween the said source and the said array are put back in phase.

[0021] Typically, for each source S_(m), the coefficients are obtainedby means of the matrix operation γ^(m)=Ω⁻¹(α^(m))* where γ_(m)=(γ₀ ^(m),. . . ,γ_(L−1) ^(m))^(T) is the vector of the coefficients for thesource S_(m), α_(m)=(α₀ ^(m), . . . ,α_(L−1) ^(m))^(T) is the vector ofthe said correlation results, L is the number of beam signals and Ω isthe matrix of size L×L whose coefficients are a function of thecorrelation of the gain functions relating to the different beams.

[0022] According to a first variant, the coefficients Ω_(l′l) areobtained by Ω_(l′l)=g_(l′) ^(H)H_(l) where g_(l) and g_(l′) are vectorsformed by angular sampling of the gain functions relating to the beamsof indices l and l′.

[0023] According to a second variant, the coefficients Ω_(l′l) areobtained by Ω_(l′l)=∫G_(l′)(θ)*G_(l)(θ)dθ where G_(l)(θ) and G_(l′)(θ)are the gain functions relating to the beams of indices l and l′respectively and where the adding is effected over the angular scanningrange defined by the array.

[0024] The invention is also defined by a device for receiving a signaltransmitted by a source or a plurality of sources comprising means forimplementing the method disclosed above.

[0025] The characteristics of the invention mentioned above, as well asothers, will emerge more clearly from a reading of the followingdescription given in relation to the accompanying figures, amongstwhich:

[0026]FIG. 1 depicts schematically a first type of known receptiondevice;

[0027]FIG. 2 depicts schematically a second type of known receptiondevice;

[0028]FIG. 3 depicts schematically a reception device according to theinvention;

[0029]FIG. 4 depicts schematically a module of the invention, accordingto a first variant embodiment;

[0030]FIG. 5 depicts schematically a module of the invention, accordingto a second variant embodiment;

[0031]FIG. 6 depicts an example of an application of the invention to aDS-CDMA system;

[0032]FIG. 7 depicts a module of the system FIG. 6 according to a firstexample embodiment;

[0033]FIG. 8 depicts a sub-module of the module of FIG. 7;

[0034]FIG. 9 depicts a module of the system of FIG. 6 according to asecond example embodiment;

[0035]FIG. 10 depicts a sub-module of the module of FIG. 9.

[0036] A first idea at the basis of the invention is based on thepossibility of decomposing a reception gain function over a set ofelementary beams.

[0037] A second idea at the basis of the invention is using thisdecomposition in order to dispense with the matrix inversion in equation(2) or (3).

[0038] The French patent application filed on 31.10.2000 in the name ofthe applicant under the number FR0014222 and incorporated here byreference shows that it is possible to decompose any gain functionobtained by beamforming (on transmission and on reception) over a set ofelementary functions. The main results disclosed therein will be set outbriefly below.

[0039] Let h be the linear application of C^(N) in the vector space F ofthe complex functions defined over the angular range in which thenetwork operates (for example [−π/2, π/2] for a linear array) whichassociates, with any vector v of C^(N), the complex gain function inreception G=h(v) obtained by weighting the antennas signal by thecoefficients v_(i), i=0, . . . ,N−1, that is to say, for example, for alinear array: $\begin{matrix}{{G(\theta)} = {\sum\limits_{i = 0}^{N - 1}{v_{i}{\exp \left( {- {{j\phi}_{i}(\theta)}} \right)}}}} & (4)\end{matrix}$

[0040] where φ_(i)(θ)=(2 πd/λ)i.sinθ is the phase difference between twoconsecutive antennas, and d and λ are respectively the pitch of thearray and the wavelength in question The image of C^(N) by h denotedIm(h) is a vectorial subspace of F with a dimension at most equal to N.Let there now be a gain function G′(θ) which it is wished to generate bymeans of the array of antennas. The function G′(θ) can be projected ontoIm(h): let G′_(P)(θ) be this projection. There exists a vector v suchthat h(v)=G′_(P) and it is possible to show that this vector makes itpossible to best approximate the gain function G′(θ).

[0041] In addition, the complex gain function G(θ) obtained (whosemodulus gives the diagram of the antenna equivalent to the array) has alimited band. It is therefore possible to angularly sample at afrequency greater than the Nyquist frequency without loss ofinformation. It is possible to show for example that the Nyquistfrequency is N rad⁻¹ for a linear array and 2N rad⁻¹ for a circulararray, which leads to taking K>πN (in practice K=4N) samples over theangular range [−π/2,π/2] in the first case and the same number ofsamples over the angular range [−π,π] in the second case.

[0042] If the samples of the complex diagram are denoted (g_(k)), k=0, .. . ,K−1, that is to say g_(k)=G(θ_(k)) where the θ_(k) values are Mangles equally distributed over [−π2,π/2] (linear array) or [−π,π](circular array) defining the sampling directions, it is possible todefine a linear application h_(s) of C^(N) in C^(K) which makes the gainvector h(v)=g=(g₀,g₁, . . . ,g_(K−1))^(T) correspond to any weightingvector v. The image of C^(N) by h_(s) is a vectorial subspace of C^(K)with a dimension at most equal to N which will be denoted Im(h_(s)). Ifa base of C_(N) is chosen, for example the canonical base and a base ofC^(K), it is possible to express the linear application h by a matrix Hof size K×N which is at most of rank N.

[0043] Let now g′ be a gain vector corresponding to any sampled complexgain function G′(θ) which it is wished to obtain. It is possible to finda vector v such that h_(s)(v) is as close as possible to g′ in the senseof the Euclidian metric on C^(K), that is to say such thath_(s)(v)=g′_(p) where g′_(p) is the orthogonal projection of the vectorg′ onto Im(h_(s)). If the matrix H is of rank N, the sought-for vector vis written:

v=H ⁺g′  (5)

[0044] where H⁺ is the pseudo-inverse matrix of H defined byH⁺=(H^(H)H)⁻¹H^(H) where H^(H) is the conjugate transpose of the matrixH. The weighting vector v will make it possible to obtain, by means ofthe beamforming, a gain function G(θ) best approximating the desiredgain function G′(θ).

[0045] In theory, N independent vectors of the space Im(h_(s)) areenough to generate it. It will however be possible to be contented, forreasons which will be seen subsequently, with generating a subspaceF_(L) of Im(h_(s)) by means of a set of independent vectors of cardinalL≦N. It will be noted that F_(N)=Im(h_(s)). The generating vectors ofF_(L) will be denoted g₀, . . . ,g_(L−1). In practice, it will bepossible to choose L=N vectors corresponding to beams pointing in thesampling directions θ_(k). The beams corresponding to the generatingvectors of F_(L) will be denoted B₀,B₁, . . . B_(L−1) and they willsimply be referred to as “generating beams”. The weighting vectorsmaking it possible to form the generating beams will be denoted v₀, . .. ,v_(L−1), or in other words h(v_(l))=g_(l).

[0046]FIG. 3 illustrates a reception device according to the invention.Beamformers 310 ₀, . . . ,3 _(L−1) can be seen therein, responsible forforming the aforementioned L generating beams. If the angular range hasa blind angle in which no signal source can be situated, no generatingbeam will be chosen in this area. The beam signals φ₀, . . . , φ_(L−1)are directed towards a combination module 315 which calculates theoutput signal y by: $\begin{matrix}{y = {\sum\limits_{l = 0}^{L - 1}{\gamma_{l}\phi_{l}}}} & (6)\end{matrix}$

[0047] where the coefficients γ_(l), l=1, . . . ,L are supplied by theoptimisation module 316.

[0048]FIG. 4 illustrates the optimisation module 316 according to afirst variant. This effects the correlation between the beam signals φ₀,. . . , φ_(L−1) and a replica of a reference signal d transmitted by thesource S, that is to say:

α_(l) =E(y _(l) d*)  (7)

[0049] The coefficients α_(l) are conjugated at 430 ₀, . . . ,430 _(L−1)in order to supply the combination coefficients γ_(l). The latter areused by the combination module 315 in order to effect a coherentcombination of the beam signals φ₀, . . . , φ_(L−1) of the MRC type. Ifthe L generating beams are separated (or in other words if the L gainvectors associated with these beams are chosen so as to be orthogonal),the reception method supplying y is optimum. On the other hand, if the Lgenerating beams are imperfectly separated, the signals of the differentbeams are correlated and the MRC combination is no longer optimum. Thisis because the interfering signals received by the imperfectly separatedbeams are also combined coherently.

[0050] If the interference level is not the same for each of the beams,the beam signals are normed prior to their combination. The outputsignal is then obtained by: $\begin{matrix}{y = {\sum\limits_{l = 0}^{L - 1}{\gamma_{l}\frac{\phi_{l}}{\phi_{l}}}}} & (8)\end{matrix}$

[0051]FIG. 5 illustrates the optimisation module 316 according to asecond variant embodiment. According to this variant, it is sought tobest approximate an optimum formation of beam by a combination ofgenerating beams.

[0052] The vector x can be expressed as a function of the signal fromthe source and a spatial transfer function:

x=as+b  (9)

[0053] where a is a vector of dimension N expressing the spatialtransfer function between the source and the array of antennas and b isthe noise vector whose components are assumed to be Gaussian.

[0054] The signal s transmitted by the source S arrives at the arraywith an angular distribution F(θ). The vector of dimension K whosecomponents are F(θ_(k)) where the angles θ_(k) define the samplingdirections will be referred to as f. This therefore gives, by linearityof the function h:

f=h(as+b)=s.h(a)+n  (10)

[0055] where n=h(b). The beam function φ corresponding to a complex gainfunction G(θ) and to an associated gain vector g can be written:

φ=g ^(T) f  (11)

[0056] The optimum vector g_(opt) consequently satisfies:

g_(opt)=(h(a))*  (12)

[0057] and hence

s*g _(opt) +n*=f*  (13)

[0058] It will be assumed initially that L=N. There is then a set ofgenerating vectors of Im(h), that is to say g⁰, . . . ,g_(N−1), on whichthe vector gpt can be decomposed: $\begin{matrix}{g_{opt} = {\sum\limits_{l = 0}^{N - 1}{\gamma_{l}g_{l}}}} & (14)\end{matrix}$

[0059] In addition, the result of the correlation of the beam signalφ_(l′) with the transmitted signal can be written:

α_(l′) =E(φ_(l′) s*)=E(f ^(T) g _(l′) s*)  (15)

[0060] and hence, taking account of (13) and (14): $\begin{matrix}\begin{matrix}{\alpha_{l} = \quad {{E\left( {\sum\limits_{l = 0}^{N - 1}{\gamma_{l}^{*}g_{l}^{H}g_{l^{\prime}}{ss}^{*}}} \right)} + {E\left( {n^{T}g_{l}s} \right)}}} \\{= \quad {{E\left( {ss}^{*} \right)} \cdot {\sum\limits_{l = 0}^{N - 1}{\gamma_{l}^{*}g_{l}^{H}g_{l^{\prime}}}}}} \\{= \quad {P_{s}{\sum\limits_{l = 0}^{N - 1}{\gamma_{l}^{*}g_{l}^{H}g_{l^{\prime}}}}}}\end{matrix} & (16)\end{matrix}$

[0061] If a normed power reference signal is used, it is possible toexpress (16) vectorially:

γ=Ω⁻¹α*  (17)

[0062] with γ=(γ₀, . . . ,γ_(N−1))^(T), α=(α₀, . . . ,α_(N−1))^(T) and$\begin{matrix}{\Omega_{ll} = {g_{l}^{H}g_{l}}} & (18)\end{matrix}$

[0063] It is reminded that the vectors g_(l) are the vectors consistingof the samples of the gain functions G_(l)(θ). Formula (18) translatesthe correlation between the gain functions of the base beams. Iffunctions G_(l)(θ) are available, this can then be written:

Ω_(l′l) =∫G _(l′)(θ)*G _(l)(θ)dθ  (19)

[0064] where the adding is carried out over the angular range of thearray.

[0065] It should be noted finally that, if the beams are decorrelated,the matrix Ω is a multiple of the identity matrix and the firstembodiment is found again.

[0066] If it assumed that L<N, it is not possible in the general case tofind a decomposition such as (14). In such instance, the vector g_(opt)is projected first onto the space F_(L) generated by the vectors g_(l).Equations (17) to (19) remain valid, the matrix Ω then being ofdimension L×L and of rank L.

[0067] According to the second variant embodiment of the module 316illustrated in FIG. 5, the beam signals φ₀, . . . ,φ_(L−1) aredecorrelated in the modules 520 ₀, . . . ,520 _(L−1) with the referencesignal d. The correlation results α₀, . . . ,α_(L−1) are then conjugatedby means of the conjugation modules 530 ₀, . . . ,530 _(L−1) beforebeing supplied to the matrix calculation module 540 calculating γ₀, . .. ,γ_(L−1) according to (17).

[0068] It should be noted that the decorrelation matrix of the beams Ω⁻¹(contrary to the decorrelation matrix of the antenna signals R_(xx)⁻¹

[0069] ) depends only on the generating beams and therefore can becalculated once and for all. In particular, it does not depend on themovement of the source S.

[0070]FIG. 6 illustrates a first example of an application of theinvention to a DS-CDMA reception system. This system is, for example,situated at a base station in order to receive the signals transmittedby a plurality M of mobile terminals, each signal having been spread bya user code C_(m). The system comprises an array of antennas 600 ₀, . .. ,600 _(N−1). The beamformers 610 ₀, . . . ,610 _(L−1) form the Lgenerating beams and produce the corresponding L beam signals φ₀, . . .,φ_(L−1). These signals are supplied to M combination modules 650 ₀, . .. , 650 _(M−1) which give the received signals z₀, . . . ,z_(M−1) comingfrom the M terminals.

[0071]FIG. 7 illustrates the structure of a combination module 650 _(m)according to a first example embodiment. In the designation of thedifferent modules, the index m has been omitted in order not to overloadthe notations. The different beam signals φ₀, . . . ,φ_(L−1) aretransmitted to L RAKE receivers 751 ₀, . . . ,751 _(L−1), whichcorrelate the beam signals with the user code C_(m) affected by thedelays of the different paths and effects the coherent combination ofthe signals issuing from the different paths. The signalsψ₀^(m), …  , ψ_(L − 1)^(m)

[0072] represent the combinations of paths within the differentgenerating beams. The module 754 provides the combination of thedifferent beams, or more precisely of the signalsψ₀^(m), …  , ψ_(L − 1)^(m),

[0073] by means of the complex coefficients γ_(l)^(m)

[0074] given by the optimisation module 755. The signal z_(m) at theoutput of the module 755 supplies an optimum estimation of the symboltransmitted by the mobile terminal S_(m).

[0075]FIG. 8 illustrates the structure of the optimisation module 755.The signals ψ₀^(m), …  , ψ_(L − 1)^(m)

[0076] are first of all correlated with the despread pilot signal D_(m)and the correlation results are then conjugated in the conjugationmodules 757 ₀, . . . ,757 _(L−1). The conjugated values thus obtainedare transmitted to a matrix calculation module effecting thecalculation:

γ^(m)=Ω⁻¹(α^(m))*  (20)

[0077] whereγ_(m) = (γ₀^(m), …  , γ_(L − 1)^(m))^(T), α_(m) = (α₀^(m), …  , α_(L − 1)^(m))^(T)

[0078] and where the coefficients Ω_(l′l) are given by (18) or (19).

[0079] The reception device illustrated by FIGS. 6 to 8 makes itpossible to best approximate an optimum reception by means of a set ofgenerating beams. Each beam can cover one or more propagation pathsissuing from a mobile terminal.

[0080]FIG. 9 illustrates the structure of a combination module 650 _(m)according to a second example embodiment. There too, in the designationof the different modules, the index m has been omitted in order not tooverload the notations. The different beam signals φ₀, . . . , φ_(L−1)are analysed by the delay analysers 953 ₀, . . . , 953 _(L−1) by meansof a correlation with the user code C_(m). Each beam signal φ_(l) isdelayed at 952 _(l) by the delays supplied by the analyser 953 _(l) andthe delayed signals are added in order to supply a signal ζ_(l). Itshould be noted that the signals ζ_(l) are aligned timewise for thedifferent paths of the user m but are not despread. The signals ζ_(l),l=0, . . . ,L−1 are then combined at 954 by means of the complexcoefficients γ_(l) ^(m) given by the optimisation module 955. The outputsignal of the module 955 is despread at 956 by means of the user codeC_(m) in order to supply an optimum estimation z_(m) of the symboltransmitted by the mobile terminal S_(m).

[0081] It should be noted that, if the paths of a user are not toodispersed in time within a beam, a more robust version without alignmentin time, that is to say without the modules 952 _(l) and 953 _(l), canbe envisaged.

[0082]FIG. 10 illustrates the structure of the optimisation module 955.Unlike the module 755, the module 955 operates before spectraldespreading. The signals ζ_(l), l=0, . . . ,L−1 are correlated at 956 ₀,. . . ,956 _(L−1) with the pilot signal dm ad the correlation resultsare then conjugated in the conjugation modules 957 ₀, . . . ,957 _(L−1).The conjugate values thus obtained are transmitted to a matrixcalculation module 958 identical to the module 758.

[0083] It is important to note that the invention can be applied to anytype of antenna array and notably to a circular array.

[0084] In addition, the reception device according to the invention hasbeen described by means of functional modules. It goes without saying,however, that these modules can implemented both by dedicated circuitsand by a processor executing all or only some of the correspondingdifferent functions.

1. Method of receiving a signal transmitted by a source by means of anarray of antennas, in which a plurality of beam signals are obtained byweighting the signals received by the different antennas, characterisedin that each of the said beam signals is correlated with a replica of areference signal transmitted by the said source and in that the saidbeam signals are combined by means of coefficients obtained by thecorrelation results.
 2. Reception method according to claim 1,characterised in that the said coefficients are obtained by complexconjugation of the said correlation results.
 3. Reception methodaccording to claim 1 or 2, characterised in that the beam signals arenormed prior to their combination.
 4. Reception method according toclaim 1, characterised in that the coefficients are obtained from alinear function of the previously conjugated correlation results. 5.Reception method according to claim 4, characterised in that thecoefficients are obtained by the matrix operation γ=Ω⁻¹α* where γ=(γ₀, .. . ,γ_(L−1))^(T) is the vector of the said coefficients, α=(α₀, . . .,α_(L−1))^(T) is the vector of the said correlation results, L is thenumber of beams and Ω is a matrix of size L×L whose coefficients are afunction of the correlation of the gain functions relating to thedifferent beams.
 6. Reception method according to claim 5, characterisedin that the coefficients Ω_(l′l) are obtained byΩ_(l^(′)l) = g_(l^(′))^(H)g_(l)

where g_(k) and g_(l′) are vectors formed by angular sampling of thegain functions relating to the beams of indices l and l′.
 7. Receptionmethod according to claim 5, characterised in that the coefficientsΩ_(l′l) are obtained by Ω_(l′l)=∫G_(l′)(θ)*G_(l)(θ)dθ where G_(l)(θ) andG_(l′)(θ) are the gain functions relating to the beams of indices l andl′ respectively and where the summing is effected over the angularscanning range defined by the array.
 8. Reception method according toone of the preceding claims, characterised in that, prior to thecombination of the said beam signals, each beam signal is subjected to aputting back in phase of the components relating to the differentpropagation paths between the said source and the said array.
 9. Methodof receiving, by means of an array of antennas, signals transmitted by aplurality of sources, in which a plurality of beam signals are obtainedby weighting the signals received by the different antennas,characterised in that, for each source, the said beam signals arecorrelated with a replica of the reference signal transmitted by thesaid source and in that the said beam signals are combined by means ofcoefficients obtained from the correlation results.
 10. Reception methodaccording to claim 9, characterised in that, for each source, prior tothe combination step and for each beam signal, the components relatingto the different propagation paths between the said source and the saidarray are put back in phase.
 11. Reception method according to claim 10,characterised in that, for each source S_(m), the coefficients areobtained by the matrix operation γ^(m)=Ω⁻¹(α^(m))* whereγ_(m) = (γ₀^(m), …  , γ_(L − 1)^(m))^(T)

is the vector of the coefficients for the source S_(m), α_(m)=(α₀ ^(m),. . . ,α_(L−1) ^(m))^(T) is the vector of the said correlation results,L is the number of beams and Ω is a matrix of size L×L whosecoefficients are a function of the correlation of the gain functionsrelating to the different beams.
 12. Reception method according to claim11, characterised in that the coefficients Ω_(l′l) are obtained byΩ_(l^(′)l) = g_(l^(′))^(H)g_(l)

where g_(l) and g_(l′) are vectors formed by angular sampling of thegain functions relating to the beams of indices l and l′.
 13. Receptionmethod according to claim 11, characterised in that the coefficientsΩ_(l′l) are obtained by Ω_(l′l)=∫G_(l′)(θ)*G_(l)(θ)dθ where G_(l)(θ) andG_(l′)(θ) are the gain functions relating to the beams of indices l andl′ respectively and where the summing is effected over the angularscanning range defined by the array.
 14. Device for receiving a signaltransmitted by a source, characterised in that it comprises means forimplementing the method according to one of claims 1 to
 8. 15. Devicefor receiving signals transmitted by a plurality of sources,characterised in that it comprises means for implementing the methodaccording to one of claims 9 to 14.